Transformer for use in a static inverter

ABSTRACT

A novel transformer is described for use in a static inverter in association with one or two switching semiconductor devices. The transformer produces an output for control of the associated switching device(s) which changes in sense from conduction aiding to conduction inhibiting as a function of the flux level in the transformer core. The invention is applicable to single loop cores, such as are assembled from two &#34;U&#34; cores. Control is effected by a primary and secondary control winding wound through an aperture pair, the aperture pair being oriented for &#34;neutrality&#34; of the second control winding to the main flux. The aperture pair creates a five branch magnetic path which permits optimizing the control voltage applied to the associated semiconductor devices both to enhance the switching efficiency when the switching device is initially turned on and to reduce stresses on the switching device by precluding transformer saturation when the switching device is turned off. With two switching devices, two aperture pairs are normally provided.

.[.This is a continuation-in-part of co-pending U.S. patent applicationSer. No. 875,337, Feb. 6, 1978, now abandoned, entitled "A StaticInverter and a Transformer for Use in a Static Inverter" by Nicholas A.Schmitz, James E. Harris and William Peil..].

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to transformers for use in static inverters.Static inverters are devices in which electrical energy in the dc formis converted to electrical energy in the ac form through static means.The present invention lies in the class of inverters in which the dcsource produces a current through one or two semiconductor devices, eachconnected in series with a primary winding of a power transformer andproducing an ac output in a transformer secondary winding as thesemiconductor devices are switched. The transformers as described hereinare adapted for use in inverters using one or two transistorconfigurations, and include control windings, which are coupled to aninput electrode of the semiconductor device to effect efficient andstress free switching.

2. Description of the Prior Art

Static inverters of the class employing a dc source, one or twoswitching semiconductor devices and a transformer are well known. Thearrangements may exist in either free running or the driven form.Representative static inverters are illustrated in U.S. application Ser.No. 956,578 of Hesler et al, now U.S. Pat. No. 4,202,031, assigned tothe assignee of the present invention. In either free running or drivenform, feedback windings may be used to effect switching of thesemiconductor devices. Recently, as represented by U.S. Pat. No.3,914,680 and U.S. Pat. No. 4,002,999 to Hesler et al and assigned tothe Assignee of the present invention, the transformer properties havebeen tailored to the requirements of the semiconductor in the inverterapplication, in particular to maximise the switching efficiency and toavoid unduly stressing the semiconductor devices resulting from fullcore saturation. In both cited patents, the main core is provided withan aperture (or two, in the two transistor inverters), each of whichdivides the core into two localized branches. One branch is designed tosaturate first, and upon saturation to reduce the regenerative andincrease the degenerative feedback applied to the transistor, so as toprevent full core saturation. The patented circuits have led to areduction in cost of such inverters, and a substantial improvement inthe switching efficiencies. The present invention represents a furtherimprovement of such arrangements.

SUMMARY OF THE INVENTION

Accordingly, it is an object of the present invention to provide animproved transformer for use in a static inverter.

It is a further object of the present invention to provide a noveltransformer for use in a static inverter in which full core saturationand the accompanying stresses on the associated semiconductor devicesare avoided.

It is still another object of the present invention to provide a noveltransformer for use in a static inverter in which the efficiency of theassociated switching semiconductor devices is improved and circuitlosses are reduced.

It is an additional object of the present invention to provide a noveltransformer for use in a static inverter in which a control output isproduced for application to a switching device which changes in sense asa function of the flux level in the transformer core.

It is still another object of the present invention to provide a novelinverter subcombination comprising a transformer and a junctiontransistor switching device interconnected to facilitate an optimumtransistor conduction control, which changes in sense as a function ofthe flux level in the transformer core and improves both inverterefficiency and reliability.

These and other objects of the invention are achieved in a transformerhaving a core of substantially linear magnetic material having a closedmagnetic path to which a pair of apertures are formed in a localizedregion of the core. One aperture divides a magnetic path into a firstand second branch, and the second aperture divides the magnetic pathinto a third and fourth branch with a fifth branch being formed betweenthe apertures. The first and fourth branches form a "first" diagonalpair of branches and the second and third branches form a "second"diagonal pair of branches. The apertures are serially arranged along themagnetic path to prevent the main flux from flowing into the fifthbranch and to place the fifth branch "orthogonal" to the main flux.

A primary winding is provided encircling the full magnetic path forgenerating a main flux in the core when current is supplied thereto. Afirst control winding is provided encircling the fifth branch forgenerating a circulating flux when current is supplied to it forming twocounter-rotating loops around the apertures. The fluxes in the two loopscombine additively in the fifth branch, with the flux in a loopcombining additively with the main flux in one diagonal pair of branchesand substractively in the other diagonal pair of branches. A secondcontrol winding is provided encircling the fifth branch.

The second control winding derives an electrical quantity whose signreverses as a function of the magnetic state of the core.

In the magnetic design, the products of the reluctances of the firstdiagonal pair of branches should be set equal to the products of thereluctances of the second diagonal pair of branches to reduce thetendency of the main flux to be coupled to the second control winding.In addition, the reluctances of the first and second branches may be setequal, a more restrictive design setting which further improves"neutrality" to main flux with nonlinear ferrites.

In a practical case, where three levels of control are sought, thereluctances of the first and second branches are set to be greater thanthe reluctances of the third and fourth branches, predisposing the firstbranch to saturate first and the fourth branch to saturate second. Thisis normally achieved by making one aperture larger than the other, andinsuring that the reluctance of the fifth branch is less than that ofthe other branches to preclude its saturation prior to the first andfourth branches.

Preferably, the first and second control windings are of a few turns andare closely coupled in the absence of selective saturation to achievesubstantial current transformer action. The magnetic design produces amaximum core coupling between the first and second control windings inthe absence of saturation, the coupling being reduced as each branchsaturates.

In an inverter, means are provided for supplying substantiallysynchronous alternating currents to the primary transformer winding andto the first control winding to insure that the main and circulatingfluxes add in the first diagonal pair of branches and subtract in thesecond diagonal pair, and force the first diagonal pair of branches tosaturate first as supplied current increases.

When a resistive load is coupled to the second control winding,saturation of the first diagonal pair of branches tends to force mainflux into the fifth branch. This causes a reversal in the direction ofthe flux coupled to the second control winding and a reversal inpolarity of the electrical control quantity coupled to the resistiveload.

When the load coupled to the second control winding is the inputjunction of a junction transistor, the mechanism of reduction andreversal in the control output involves the repeated increase in thereluctance of the current transformer formed by the first and secondcontrol windings.

Typically, the junction transistor is the switching device for supplyingalternating current to the primary winding and to the first controlwinding and the device exhibits appreciable stored charge. When thesecond winding is coupled across the input junction of the transistor,current transformer action from the first control winding tends to applya conduction aiding base current to the transistor at the onset ofconduction. A property of the connection of the input junction acrossthe second control winding is that a constant voltage drop is sustainedacross the second control winding so long as the transistor remainsconductive. This fixed voltage in turn imposes a constant rate of changeof flux on the magnetic region (the fifth branch particularly) to whichthe second control winding is coupled. Accordingly, when the one branch(the first) saturates, assuming unequal apertures, the reluctancescoupling the first and second control windings together is increasedappreciably and reduces the rate of increase of applied base current.When the next branch (the fourth) saturates, the reluctance is increasedso substantially that both a slope reversal and an absolute currentreversal is produced. The reversal in current drive continues untilstorage charge is removed from the switching device and its conductionterminated.

The arrangement described herein achieves high efficiency switchingthrough the provision of a strong regenerative drive during the initialconduction interval followed by an optimum transition to the OFF statewith controlled reversal of current drive. The timing and rate ofreversal can be adjusted to prevent full core saturation and theresultant stressing of the transistor.

The arrangement is applicable to inverters using one or two switchingdevices. With two switching devices, two aperture pairs may be used.

BRIEF DESCRIPTION OF THE DRAWING

The novel and distinctive features of the invention are set forth in theclaims appended to the present application. The invention itself,however, together with further objects and advantages thereof may bestbe understood by reference to the following description and accompanyingdrawings, in which:

FIG. 1 is an electrical circuit diagram of a static inverter employing anovel power transformer having control windings which in response to themagnetic state of the core provide successive conduction aiding andconduction inhibiting feedback;

FIG. 2 is a drawing illustrating the construction of the noveltransformer including a core having unique aperture pairs and thewindings associated with the core and the aperture pairs;

FIGS. 3A through 3D are explanatory diagrams of the magnetic state ofthe transformer core in the region of the aperture pair at successivestages in the transistor switching cycle;

FIGS. 4A through 4C are schematic diagrams useful in a mathematicalanalysis of the transformer control function;

FIG. 5 are three idealized waveshapes predicted by the analysis andrepresentative of the device operation; and

FIG. 6 is an electrical diagram of an inverter employing two alternatelyconducting transistors.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to FIG. 1, a dc to ac inverter employing a noveltransformer is shown for converting electrical energy supplied from a155 V dc source 11 to a load at 200 volts at approximately 25 kilohertz.The ac load may be a high efficiency gas discharge lamp as illustratedat 12. The inverter comprises a power transistor 13, the novel powertransformer 14, sundry circuit elements 33, 34, 35 associated withtransistor 13 and a trigger oscillator 24, the combination (less 24)functioning as a forward converter in which energy during the off stateof the switching transistor is returned to the dc source 11.

The novel power transformer 14, illustrated in greater detail in FIG. 2,has a single rectangular loop core 15, a primary winding 16, a forwardconversion winding 17, and a secondary winding 18, all coupled about thefull core cross section. The 126 turn primary winding 16 and the 126turn forward conversion winding 17 are bifilarly wound about the leftleg of the core for close coupling as illustrated in FIG. 2. Inaddition, 70 turns of the 215 turn secondary winding are wound on theleft leg of the core for increased coupling to the primary, the 145 turnremainder being wound around the right leg of the core. The powertransformer also has a primary 19 and a secondary 20 control winding forconnection to the power transistor 13. These control windings are ledthrough a first double aperture (36,37), which as will be explained,produce a mutual coupling dependent upon flux levels in the core. Asecond double aperture (38, 39) with which windings 21 and 22 areassociated is provided for operation of the trigger oscillator 24. Thetrigger oscillator output appears at a third control winding 23 woundthrough the first double aperture (36, 37) for coupling to the powertransistor input, as will be described in greater detail hereinafter.

The transistor inverter employs the transformer 14 and the switchingtransistor 13 in a triggered forward converter configuration. Theswitching transistor is turned on by a trigger oscillator 24 timed toproduce a pulse at a 25 KHz repetition rate, and the magnetics determinethe shape of the conduction waveform and the length of the conductionpulse. To achieve that control, the control windings initially provide aregenerative or conduction aiding drive followed by a degenerative orconduction inhibiting drive which effects transistor turn off.

The trigger oscillator 24, which is not in itself a part of the presentinvention, is similar to that described in U.S. application Ser. No.974,253, filed Dec. 28, 1978 by William Peil, entitled "A PulseGenerator Producing Short Duration High Current Pulses for Applicationto a Low Impedance Load" and assigned to the Assignee of the presentapplication. The trigger oscillator, which utilizes the magneticstructure of the power transformer 14 as noted earlier, comprises thetransistor 25, resistors 26, 27, 28, capacitors 29, 30, the diode 31 andthe transformer windings 21, 22 associated with aperture pair 38, 39 andwinding 23 associated with aperture pair 36, 37.

The trigger oscillator 24 is a relaxation oscillator having magneticallycoupled regenerative feedback essential to the production of a highintensity short duration trigger pulse and a biasing configuration whichmakes the pulse repetition rate insensitive to variations in the dcsupply voltage or load.

The trigger oscillator consists of an NPN transistor 25 having itscollector electrode connected through the trigger output winding 23 tothe positive terminal 32 of the dc source and its base electrodeconnected through a protective diode 31 to a voltage divider consistingof the resistances 26 and 27 connected in the order recited between theB+ terminal 32 and ground. The emitter of transistor 25 is connected toground through the serially connected primary feedback winding 22 andthe resistance 28. A capacitor 30 is provided coupled between theungrounded terminal of the resistance 28 and the B+ terminal 32.Secondary feedback winding 21 is coupled by capacitance 29 across theserially connected diode 31 and the input junction of transistor 25. Thediode 31 and transistor input junction are connected in like polarity.

The trigger oscillator functions as a relaxation oscillator with thecapacitor 30 being recurrently charged through the resistor 28 andrecurrently discharged through transistor 25. In the charge-dischargeprocess, the voltage on the lower terminal of the capacitor falls slowlyfrom near B+ to a value of typically 15-40 volts below B+ at a dischargerate established by resistor 28, the size of the capacitor and the B+potential. At the desired minimum voltage, the transistor becomesconductive, arresting the downward discharge. Current flow through thetransistor 25 occurs between the upper B+ connected capacitor terminaland the other, lower capacitor terminal, transistor conduction bringingthe lower capacitor terminal to a potential slightly (i.e. 2 volts)below B+. When the discharge stops with the transistor input junctionstrongly back-biased, the charging through resistance 28 repeats.

The charging of the capacitor is arrested when the transistor 25 becomesconductive at a voltage set by the base connected voltage divider. Theemitter electrode, which is connected through the low impedance winding22 to the lower capacitor terminal, follows the potential of the lowercapacitor terminal as it falls. The base electrode of transistor 25,however, which is connected through diode 31 to the voltage divider 26,27 connected across the dc source, is maintained at an arbitraryfraction of the B+ potential (about 15-40 volts below B+). Thus, thetransistor input junction varies from a strong (15-40 volts) backwardbias precluding conduction when the capacitor begins to charge, to aforward bias causing the transistor to become conductive again.Transistor conduction halts the charging process with an abruptdischarge of the capacitor. Transistor conduction begins when the lowerterminal of the capacitor 30 is approximately two diode drops below thevoltage of the base connected tap on the voltage divider.

The transistor turn on mechanism is affected by the parasitics of thetransistor and the shunting effect of the low impedance feedbackwindings. At near zero base current, the base impedance is high and thetransistor ac gain is low, attributable to a dc β roll off, and to avery low high frequency cut-off due to parasitic capacity shunting theinput and output terminals. Thus, the initial onset of base currentconduction will not produce a greater than unity current gain condition.As the base current continues to increase, however, the base inputimpedance will fall and the high frequency cut off will increase,providing increasing ac gain. When the output current reaches a valuewhere the shunting effect of the low inductive reactances of the primaryand secondary feedback windings no longer keep the circuit gain belowunity, effective regenerative action will occur.

With effective regeneration provided by the feedback windings 21, 22,full transistor conduction takes place very quickly. Current flowsthrough the winding 23, the transistor 25, and the winding 22 in aclosed path carrying current from the upper to the lower terminal of thecapacitor 30. The current flowing in the primary feedback winding 22induces a regeneratively sensed base drive in the secondary feedbackwinding 21 magnetically coupled to the core through apertures 38, 39.The feedback causes a very sudden increase in current in the transistor,permitting the capacitor to discharge quickly. The discharge throughoutput winding 23 wound through aperture pair 36, 37 induces a pulse offrom 0.5 to 1 ampere having a duration of approximately 200 nanosecondsin the second control winding 20 of the main switching transistor 13turning it on and starting the conduction cycle. In the example, thepulse repetition rate is 25 KHz.

The aperture pair 38, 39 by means of which the feedback windings 21 and22 are inductively coupled, is positioned in the core for neutrality tothe main flux in the same manner as the aperture pair 36, 37 used forcontrol of the switching transistor 13. The trigger pulse is generatedby the trigger oscillator at a time when the switching transistor 13 isquiescent. Conversely, once the switching transistor is conducting, thetrigger oscillator is quiescent. Both factors reduce the chance foradverse interaction. The magnetic isolation attributable to the geometrybetween the main flux and the trigger oscillator feedback windings 21and 22 is generally sufficient for isolation throughout the switchingcycle.

The trigger oscillator described has a repetition rate which issubstantially independent of variations in source voltage or in loading.

The forward converter is not in itself a part of the present invention,and is similar to that described in U.S. application Ser. No. 956,578filed Nov. 1, 1978, now U.S. Pat. No. 4,202,031, and assigned to theassignee of the present application.

The triggered forward converter consists of the switching transistor 13,the power transformer 14, including windings 16-20, 23, diodes 33, 34and resistance 35. These components are interconnected as follows. Theprimary power winding 16 and the primary control winding 19 are seriallyconnected between the B+ terminal 32 and the collector of transistor 13.The emitter of transistor 13 is connected to ground. A secondary controlwinding 20 is coupled between base and emitter of transistor 13, with aresistance 35 connected in shunt with the input junction to prevent freerunning. A decommutating diode 33 is coupled between the connection ofwindings 16 and 19 and ground in a polarity to reduce charge storage inthe transistor 13 during current reversal. The forward conversionwinding 17, closely coupled to the primary winding 16, has its undottedterminal coupled to B+ terminal 32 and its dotted terminal coupledthrough diode 34 to ground. Diode 34 is poled to re-inject energy storedin the magnetics into the power supply during transistor turn off, afeature which also reduces stresses on the transistor. The secondarywinding 18 of which 70 turns are closely coupled to the primary winding,drives the load 12, providing a 200 V p to p voltage for a gas dischargelamp load at approximately 25 KHz. The control windings 23, 19 and 20supply the initial ignition pulse to the switching transistor, andsupply an optimized drive to the transistor input electrodes inaccordance with the invention.

The forward converter operates in the following manner. The switchingtransistor 13 is initially turned on by a high current short durationpulse inductively coupled from the output winding 23 of the triggeroscillator to the winding 20, coupled to the base of transistor 13. Thepulse causes the transistor 13 to start to conduct with collectorcurrent flowing in the primary feedback winding 19. Once started, thecollector current waveform increases in a substantially linear upwardramp. The ramp extends through the conduction interval at a slopedetermined by the primary inductance of the transformer 14. After thetriggering instant, the base current induced in winding 20 firstsustains transistor conduction, and then reversing, terminatestransistor conduction. Intermittent conduction by the switchingtransistor 13 causes an alternating current in the primary winding 16 ofthe power transformer, inducing the alternating voltage in the secondarywinding 18 suitable for operation of the load 12.

Waveforms useful to understanding the operation of the forward converterare illustrated in FIG. 5. The output voltage of the inverter (notshown) is a substantially rectangular waveform of slightly under 200 Vpeak to peak (under no load) at a 25 KHz repetition rate. The amplitudeand shape of the waveform is strongly dependent upon the load and isnormally of lesser amplitude and lesser rectangularity in the loadedcondition. The collector current waveform is the lowermost waveform ofFIG. 5. As indicated above, it is a linear ramp during the conductioninterval, falling sharply and remaining at a zero value until the nextconduction interval. The base current waveform, second from the bottomin FIG. 5, illustrates the desired initial steep current increase, theless steep portion, followed by slope reversal, and then drive reversalwhich terminates transistor conduction. The base emitter voltage is theuppermost waveform in FIG. 5. Throughout transistor conduction, thebase-emitter voltage (V_(be)) holds at a nearly fixed forward value(+0.7 volts). At the instant of collector current collapse (after storedcharge removal), the base voltage goes strongly negative, and thenreturns to zero until the next trigger pulse.

The base drive waveform (I_(b)) illustrated in FIG. 5 and supplementedby charge stored from the trigger pulse, sustains transistor conductionimmediately after the termination of the trigger pulse. Later in thecycle, the waveform exhibits a current feedback characteristicparticularly desirable when one wishes to turn off a power transistor ina typical inverter circuit. The characteristic permits one to turn thetransistor off at the end of each conduction cycle in advance of fullsaturation of the inverter transformer. Should saturation not beprevented, the inductive impedance in the transistor load will fallsharply, applying the full B+ potential across the transistor andsubjecting it to very high current stresses. In addition to preventingturn off stresses, the base drive should be made strongly regenerativeat the initial portion of the cycle in the interests of optimumswitching efficiency.

In accordance with the invention, the optimized base drive justdescribed is achieved by means of the control windings 19, 20 and 23associated with apertures 36 and 37 in the transformer core.

FIG. 2, which shows the ferrite core 15, the disposition of the powerwindings and the control windings also illustrates exemplary coredimensions. As earlier indicated, the primary winding 16, the forwardconversion winding 17 and 70 turns of the secondary winding 18 embracethe left leg of the full core cross section with the remainder of thesecondary winding 18 being wound on the right leg. The upper, lower,right, and left legs have a 0.38"×0.38" square cross section. The corehas outside dimensions of approximately 11/2"×11/2×0.38" and it isassembled from two "U" cores with a 0.020" air gap at the joints. Theaperture pair 36, 37, with which the control windings 19, 20 and 23 areassociated is disposed in a region in the upper left corner of the corewhile the aperture pair 38, 39 with which the windings 21, 22 areassociated is similarly disposed in a region of the lower left corner ofthe core near the primary bobbin. In each case, the smaller apperturehas a diameter of 0.06" and is disposed on the center line of itsrespective upper or lower leg, leaving a 0.160 margin above and belowthe aperture. The larger aperture, in each case, is 0.100" in diameterand is disposed along a diagonal line extending through the corners ofthe core. The distance between the large aperture and the inside cornerof the core is 0.140" thus being 0.020" less than the 0.160" wall towall spacing of the smaller aperture. The interaperture distance is0.190", being selected to be larger than either the 0.16" or the 0.14"dimensions.

The two apertures 36, 37 whose positions and dimensions have just beendiscussed divide the main flux path into five branches. These branchesare illustrated in each of FIGS. 3A through 3D under differing magneticstates. As seen in FIG. 3A, the core portion above the small aperture 36is assigned a reference numeral 40, while the portion beneath theaperture is assigned a reference numeral 41. The core portion betweenthe outer extremity of the core and the large aperture 37 is assigned areference numeral 43, while the portion between the inside corner of thecore and the aperture 37 is assigned a reference numeral 44. The coreportion between the apertures is assigned a reference numeral 42.

The overall objective of providing an optimized base drive waveform forthe switching transistor which is first strongly conduction aiding andfinally conduction inhibiting is achieved by several magnetic designfeatures.

The first feature is that of orthogonality or "neutrality" in thedisposition of the inter-aperture branch 42 and the associated secondcontrol winding with respect to main flux (φ_(m)) flowing around themain flux path of the core. If one assumes that no main flux is flowing,and that the only flux is the circulating flux (φ_(c)), generated by theprimary feedback winding 19, one will find that the indicateddisposition produces closely coupled current transformer action betweenthe primary and secondary control windings 19 and 20. As the current inthe collector winding increases, then a proportional increase will occurin the base connected winding, and if the windings 19 and 20 areproperly sensed, a conduction aiding base current will be created,tending to be proportional to the turns ratio. In the present example,the collector winding is one turn and the base connected winding is ofthree turns, tending to force the base current above the natural betadetermined current, and further enhancing the drive. "Neutrality" in thesense used here means that if one assumes that current flows in only themain windings 16 and 17 creating a flux (φ_(m)) in the main flux path,that there will be substantially no drive induced in the secondarycontrol winding 20.

The "neutrality" to main flux of the base winding, as just described, isachieved by orienting the two apertures (36, 37) downstream of oneanother in the main flux path so that no flux tends to flow into theinter-aperture region (42). Had the two apertures been located on a legremote from the corners, the "neutral" position for the two apertureswould be in a line approximately parallel to the center line of the leg(and normally on the center line). In practice, it is preferable toplace the apertures in close proximity to the main power winding toreduce the time delays in the switching device. In addition, the largeraperture should be placed in the corner so that neither the total crosssection of the core nor the strength of the core is reduced by itspresence.

With the foregoing geometry and depending in part upon the nature of theload applied to the second control winding 20, it will respond only tothe circulating flux generated by the control winding 19 and will notrespond to the main flux (φ_(m)) until the main flux has achieved quitehigh levels. More particularly, in the event that the load coupled tothe control winding is the input junction of a transistor havingappreciable stored charge, it has been found that any diversion of mainflux into the inter-aperture branch (42) will be resisted until thetransistor becomes non-conductive. In the event of a resistive loadlacking energy storage, the inter-aperture branch 42 may still retainsubstantial neutrality to the main flux until one or two of the branches(40, 44) are saturated. The initial neutrality to main flux justdescribed under both load conditions permits one to achieve an initiallystrong conduction aiding drive as a result of simple current transformeraction between windings 19 and 20, uninterferred with by main flux.

A second feature in the magnetic design produces the reduction inforward drive and then reversal. This feature is the proper dimensioningof the cross sections of the five branches and thereby theirreluctances, as set forth earlier. As will be shown, if one branchadjacent to the larger aperture (e.g. 44) is dimensioned to saturatefirst, and the branch adjacent the smaller aperture 36 (e.g. 40) isdimensioned to saturate next, the inter-aperture branch 42 normallybeing dimensioned not to saturate, and the diagonal reluctance productsare equal (R43×R41=R40×R44), then the desired reduction in drive andfinal drive reversal will occur (as will be explained below).

The desired reversal in drive may be achieved by one of two mechanismsdependent on the load coupled to the secondary control winding. In thecase where the principal load to the winding 20 is the input junction ofthe switching transistor 13 which exhibits appreciable stored charge,base drive reduction and reversal may be produced by a change in thereluctance coupling the primary and secondary control windings 19, 20 asthe branches 44 and 40 saturate. The presence of stored charge in theinput junction, as will be explained, tends to sustain a constant rateof change of flux in the inter-aperture branch (φ) until the charge isswept out. Under this condition, substantial neutrality to the main fluxis sustained throughout the base drive cycle in spite of localizedsaturation.

In the event that the load coupled to the control winding 20 is a simpleresistance, not having an energy storage property, then main flux mayenter the inter-aperture branch 42 flowing from branch 43 to 41 asbranches 44 and 40 become saturated, and produce a reversal in drive bya second flux steering mechanism. The new diagonal flow of the main fluxinduces a current in winding 20 in a sense reversed to that generated bythe primary winding 19.

These matters will now be undertaken in greater detail with reference toFIGS. 3A through 3D.

FIG. 3A is a view of the initial flux conditions in the vicinity of thedouble aperture. Solid arrows depict the flux conditions a moment innormal operation after the trigger pulse, and transistor currents havebegun to flow in the control windings 19 and 20. The main flux φ_(m)enters from below the corner of the core and exits to the right of thecorner pursuing a clockwise course around the core above and below theaperture pair. In the vicinity of the apertures 36 and 37, a circulatingflux (φ_(c)) is created around each aperture as a result of currentflowing in the control winding 19. The serially connected controlwinding 19 carries the same current that flows through primary winding16, which generates the main flux φ_(m), fixing the phase relationshipsbetween φ_(m) and φ_(c). Typically, the circulating flux attributable tothe control winding 19 is made counterclockwise around the aperture 37and clockwise around the aperture 36 when the main flux has a clockwisesense around the main core.

The flux distribution near the apertures (36,37) may be regarded asresulting from a pair of magnetomotive forces generating main andcirculating fluxes in a five branched magnetic path.

Considering one branch at a time, the main and circulating fluxes inregion 44 adjacent aperture 37 are in the same direction and thus add,while in the branch 43, they are in an opposite direction and subtract.Thus, as between branches 44 and 43, branch 44 has the highest fluxlevel, and if the relative cross-sections and path lengths leave thereluctance of branch 44 equal to branch 43, one may expect branch 44 tosaturate first--should current in the main and primary control windingcontinue to increase. Similarly in respect to the regions adjacentaperture 36, the main and circulating fluxes add in region 40 andsubtract in 41, leading to a greater flux in branch 40 than in 41.Should current in the main and primary control windings continue toincrease, and if both regions are of equal path length and crosssection, one would expect branch 40 to saturate before branch 41. Shouldall four branches (40, 41, 43, 44) be considered together, then due tothe lesser cross-section of branch 44 (0.140×0.380) in respect to thecross-section of 40 (0.160×0.380), and assuming equality in the totalflux in the two branches, one would expect the branch 44 to saturate inadvance of the branch 40.

The dimensions of the apertures have been selected to insure theforegoing saturation sequence. FIG. 3B illustrates the second state withbranch 44 saturated and symbolized by shading lines. The inner main fluxinside the aperture pair is shown dashed to symbolize passage through asaturated branch tantamount to an air gap. The flux circulating aboutaperture 37 (now shown dashed) must now pass through a saturated branchalso tantamount to an air gap. The magnetic coupling between primary andsecondary control windings is substantially halved, since the left handmagnetic toroid encircling the control windings (19,20) as they passdown through the core is destroyed, leaving only the right hand magnetictoroid encircling the control windings (19,20) as they pass up throughthe core. Calculation also shows that the core reluctance coupling theprimary to the secondary control winding, should increase by a factor ofapproximately two. This accounts for the initial decrease in the slopeof the base current i_(b) shown in FIG. 5.

FIG. 3C illustrates the third state with shading lines symbolizingsaturation of the second branch 40. Both main flux paths above and belowthe aperture pairs are shown dashed, implying passage through an airgap. The flux circulating about both apertures 37 and 36 are bothdashed, since both portions must now pass through a saturated branchtantamount to an air gap. When branch 40 saturates, the right handmagnetic toroid encircling the control windings 19, 20, as they returnupward through the core is also destroyed. Thus, the close magneticcoupling between primary and secondary winding (19, 20) is almost fullydestroyed. Now all magnetic paths associated with windings 19, 20 havehigh reluctances. That is, two saturated paths exist, one around eachaperture and a third longer path exists around the main core threadingthrough branches 41, 42 and 43, including the two 0.020" air gapsbetween the core halves. Calculations show that these three paths are ofhigh reluctance relative to the original double toroid by two orders ofmagnitude, typically. The reluctance increase accounts for the verysudden reversal in slope of the base drive current, i_(b), (FIG. 5). Theslope of the base drive continues downward through an inversion in thesense of the current drive (i_(b)), extracting stored charge from thetransistor input junction and turning if off completely. The junctionvoltage remains positive (+0.7 volts) until the last of the storedcharge is removed, and then goes strongly negative (as shown in FIG. 5)returning to near zero until the next trigger pulse causes conduction tostart again.

Prior to a more analytical treatment of the foregoing drive reversalmechanism, a second drive reversal mechanism, which occurs with anon-energy storing load coupled to the control winding 20, will bedescribed. Under these load conditions, saturation of branch 44 occursfirst (as before) and incremental main flux is forced into branch 43. Atthis point the main flux increments may continue to flow through branch40 with some diversion into the inter-aperture branch 42. As circulatingflux and main flux continue to grow, branch 40 where the main flux andcirculating flux add, now saturates. Saturation of branches 40 and 44practically destroys current transformer action between 19 and 20 asbefore, also reducing any incremental growth in the circulating flux(φ_(c)) in the inter-aperture branch 42 to a negligibly small value. Theincremental main flux now substantially barred by saturation fromentering branches 44 and 40, increases in branches 43 and 41, and passesstrongly through the inter-aperture branch 42. The sense of the mainflux in the branches 43, 42, 41 is opposed to that produced by thecirculating flux, and since the main flux is larger, a strong conductioninhibiting voltage is applied to the secondary control winding 20. Theforegoing mechanism applies if a large resistance is placed in serieswith the input junction of a transistor weakening the fixed V_(be)voltage and energy storage constraints. The waveshape with eitherloading, however, exhibits as initially strong conduction aiding drive,a reduction in forward drive, followed by a strongly conductioninhibiting drive.

A more exact understanding of the operation of the invention may beachieved by mathematical analysis. FIG. 4A illustrates a two aperturearrangement generally similar to that illustrated in FIG. 2 butdiffering in several respects. The aperture pair is illustrated disposedalong the center line of a straight magnetic path, assumed to be part ofa closed magnetic loop. The main flux is shown entering on the righthand side and exiting on the left hand side. The circulating fluxesassociated with the apertures are shown rotating clockwise around thelarger aperture and counterclockwise about the smaller aperture.Symbolic reluctances are shown positioned in each of the five branches40-44 with subscripts assigned to designate each branch. In FIG. 4B thereluctances are interconnected into a magnetic equivalent circuit. Forthe analysis, three flux loops (φ₁, φ₂, φ_(m)) have been defined. Thefirst flux loop (φ₁) embraces R40, R42, the generator U_(h) and R41.U_(h) is the magnetomotive force acting on the inter-aperture branch 42provided by the primary control winding 19 and the secondary controlwinding 20. The second flux loop (φ₂) embraces R40, R43, R44 and R41.The third loop (φ_(m)) embraces the main flux generator φ_(m), R41 andR44. From inspection, the flux in the inter-aperture branch φ₄₂ (now φ₁)is defined by the following expression: ##EQU1## The coefficient of thenumerator term "φ_(m) " may be symbolized by "Δ_(N) ": ##EQU2## Thedenominator may be symbolized as "Δ_(D) "; ##EQU3## Assuming linerreluctances before saturation and symmetry in aperture placement bywhich R41 R43=R40 R44, the numerator term Δ_(n) becomes zero. Thus,prior to saturation, φ₁ is independent of main flux to the extent thatΔ_(N) approaches zero.

The virtual generator resulting from current flow in the controlwindings 19 and 20 produces a magnetomotive force U_(h) :

    U.sub.h =N.sub.c i.sub.c -N.sub.b i.sub.b                  (4)

Expression 4 is a consequence of Ampere's law and Lenz' law, the latterlaw implying that induced current flowing in the secondary winding N_(b)(winding 20) generates a magnetomotive force opposing the primarycurrent flowing in the primary winding N_(c) (winding 19).

Continuing the analysis, the simplified electrical circuit involving theprimary winding N_(c) (winding 19) and the secondary winding N_(b)(winding 20) are illustrated in FIG. 4C. A transistor is illustrated inthis figure, having its collector coupled to a suitable source of B+potentials, its base led through an external resistance R_(b) and thesecondary winding N_(b) to its emitter. The emitter is then connectedthrough the primary winding N_(c) to ground. An arrow indicating thebase current (i_(b)) is shown flowing through the winding N_(b) and theresistance R_(b) toward the base. The collector current i_(c) is shownflowing through the winding N_(c) to ground. A voltage V_(be) isindicated between the base and emitter electrodes. The quantity V_(be)is assumed to be constant in the transistor in the forward biaseddirection and to remain constant until all stored charge is removed.

By Faraday's law, the voltage induced in the secondary winding mustequal the voltage drop in the external base circuit:

    N.sub.b φ.sub.1 =i.sub.b R.sub.b +V.sub.be             (5)

The quantity R_(b) is normally kept quite small and is zero in thepresent practical circuit.

As a further simplification in the analysis, the main flux (φ_(m)) andcollector current (i_(c)) are assumed to be a linear function of time:##EQU4## Where L_(p) is the inductance of the primary winding and N_(p)are the turns in the primary winding. These assumptions are consistentwith observed collector current waveforms. Combining the previousequations, a differential equation may be written which specifies theflux (φ₁) in the inter-aperture branch. ##EQU5## When this expression issolved by use of the La Place transformation, the following expressionis obtained: ##EQU6## Using expressions 1 and 4 and solving for i_(b) :##EQU7## Using expression 6 to eliminate φ_(m) : ##EQU8## Assuming from(6) that i_(c) (t)=B+L/_(p) t, i.e. a linear(11) function of timethroughout the conduction period, i_(c) may be eliminated: ##EQU9##Substituting for φ₁ from expression 8 and simplifying we obtain:##EQU10## Assuming that the base time constant (N_(b) ² /R_(b) Δ_(D)) isseveral times larger than the time constant required to saturate theferrite and establish transistor turn off, expression 13 can be furthersimplified: ##EQU11##

The terms of expression 14 have several implications. The bracketedquantity is multiplied by the quantity t, making it clear that i_(b) isa time dependent quantity. The first term in the bracket corresponds tothe current transformer and signifies the contribution to i_(b)resulting from the windings 19 and 20, the B+ potential and theinductance of the main primary winding. The second term of expression 14contains the numerator term Δ_(N), which is initially zero, assuminggeometrical symmetry and nonsaturation. The last term is a virtualcurrent generator term reflecting the presence of the transistorjunction. The second term will take the non-zero values when saturationof branches 40, 44 occur and will serve to reverse the base drive sincethe quantity Δ_(N) will assume large negative values.

Expressed mathematically, a neutral flux disposition is achieved byplacing the large aperture 37 and the small aperture 36 along thepractical center line of the leg, so that the reluctances in thebranches above and below each aperture are alike, while the reluctancesassociated with the larger aperture are larger than the reluctancesassociated with the smaller aperture. The equations support the moregeneral concept that the diagonal products of the reluctances should beequal (R41×R43=R40×R44). This implies that there is no immediate loss inisolation if the apertures are displaced from the center line, so longas the centers of the apertures are held parallel to the center line.The core material is not perfectly linear, and tends to imbalance therelationship as the offset increases, however. The reluctance of theouter and inner paths is altered, at the corners attributable in part tothe greater length of the outer path than the inner path. The flux willtend not to enter the outer corner and individual lines of flux willturn with an appreciable radius. The composite effect is to bring theaperture at the corner in slightly for neutrality assuming a rectangularinner and a rectangular outer corner. In general, where minimum coreweights are sought and practical low cost core materials must be used,the preference is to place the apertures so that the reluctances at eachaperture are approximately equal. The aperture separation shouldnormally exceed that in the lateral branches to avoid saturation in thecenter region which also tends to make core material usage inefficient.In the practical application herein described, saturation in the centeris normally avoided by a small excess in cross section (e.g. 0.19 vs0.16 vs 0.14). The aperture separation should normally be less than twotimes the width of the core (0.38 in the present case) since thispermits two complete minimum reluctance toroids to be erected aroundeach aperture. For greater separations between apertures, the neutralityof the control winding to main flux levels suffers.

While the double aperture configuration has been employed in an inverterusing a single transistor, it may also be employed in an inverteremploying two alternately conducting transistors as shown in FIG. 6.

In FIG. 6 the transistors 51 and 52 are the main switching transistors,and the transistor 53 is a starting transistor. The power transformer 54has a center tapped primary winding and a main secondary winding coupledto the load illustrated as a gas discharge lamp 55. The core oftransformer 54 is shunted .[.as illustrated in Figure,.]. and has twopairs of double apertures, one pair in the upper right side of the coreand the other pair in the lower right side of the core using theorientations of FIG. 2. The transformer has a pair of voltage feedbackwindings 57 and 58 each connected in series with a small resistanceacross the input junctions of the switching transistors. Currentfeedback winding pairs 59, 60 and 61, 62 associated with doubleapertures are also provided. The primary windings of the currentfeedback windings are connected in series with the respective halves ofthe center tapped transformer primary and the emitters of the associatedpower transistor. This connection forces the primary current to passthrough the current feedback windings 59 and 61. The second currentfeedback windings 60 and 62, which are inductively coupled to thecontrol windings 59 and 61, respectively, are each connected with aserial resistance between the emitters and bases of the respectiveswitching transistors. Once oscillation is instituted, the voltage andcurrent feedback produce an alternating switching sequence, and the basecurrent drive applied to the individual switching transistors isinitially conduction aiding and then conduction inhibiting as before.

The FIG. 6 circuit commences to oscillate following a starting pulsegenerated by a circuit consisting of transistor 53, associated resistorsand capacitors and windings 63 and 64 associated with one aperture pair.The starting pulse appears in winding 64 and is coupled to the currentfeedback winding in the same aperture pair. The appropriate switchingtransistors (51 or 52) begins to conduct as a result. Selectivesaturation of the magnetic domains (as previously detailed) determinesthe length of the conduction interval. The turn-off induces an equalmagnitude current, but in the opposite direction in the other half ofthe primary. This current is conducted partially by the flyback diodes,each shown connected between an outer primary terminal and the collectorof one of the switching transistors. The remaining oppositely directedcurrent serves to inject charge into the base and results in base tocollector reverse conduction. The base potential soon falls below thatof the collector, and the transistor conducts normally, setting up theconditions for repeated alternate conductions. During each conductionperiod, base current is provided via current feedback to insuretransistor saturation. For certain loads (resistive loads), commutationconditions are insured without the use of flyback diodes. The reactivecurrent for these loads may be carried without harm by thecollector-base junctions.

The voltage feedback provided by winding 57 and 58 is to insure theoperation of the inverter under no load conditions. It is possible toprovide no load operation by other means such as a large magnetizingcurrent but voltage feedback is preferable since it keeps the current inthe transistor to a minimum. The illustrated circuit is suitable for usewith a number of loads including fluorescent lights. If used with agaseous discharge light source, means for supplying a high voltagestarting voltage should be provided.

Both the one transistor and the two transistor inverter power circuitswhich have been described exhibit excellent efficiencies. Knowninverters operate with 6 to 8 percent losses. With the use of the twoaperture core configurations in which drive is optimized, the systemlosses are typically reduced a further 1 to 3 percent. Since thetransistor losses contribute about one third to the net system losses,the reduction in transistor losses may be in excess of 50%, greatlyreducing the internal dissipation.

The transistor control mechanism of the present novel transformerpermits lower total circuit dissipation than certain of the citedmagnetic state responsive circuits. When a transistor input junctionbecomes the load to the control winding in the present double apertureconfiguration, the input junction may be directly coupled across thecontrol winding without the addition of a series voltage droppingresistance to protect the input junction during turn-off and establishthe reverse current level.

A resistance for junction protection with its attendant losses is notrequired in circuits coupled to the present transformer. In both theconduction aiding and conducting inhibiting modes of operation and untilthe transistor switch is turned off, the virtual electric generatoracting through the second control winding 20 (FIG. 1) remains a currentsource not capable of generating a stiff inverse voltage across thetransistor input junction. After the first branch (44) has saturated,which reduces the forward drive, the second control winding continues tomaintain current transformer action. In this state, coupling of thecontrol winding 20 to the primary control winding continues even thoughreduced and coupling of the control winding to the main flux is affectedby the presence of two alternate paths for the main flux. One flux path(40) is not substantially coupled to the second control winding and onepath (42) is. The uncoupled path (40) shunts the other path (42) andprovides an alternate low reluctance path so that main flux is notforced into branch 42 and the "stiff" voltage transformer actionmentioned earlier is avoided. In addition, the flux in branch 42 remainsload responsive, being affected by the presence of the input junction,which acting as a generator substains the existent rate of change influx in the branch and resists entry of the reversely sensed main flux.Since the ferrite in branch 40 has some residual slope in its B/Hcharacteristic, the load impedance presented to the input junctioncontinues to remain large in relation to the small internal resistanceof the junction. In consequence, the alternate flux path presented bybranch 40 remains available to (and favored for) the main flux and thetransformer retains its nature as a current source until the negativedrive has removed all stored charge from the transistor input junction,and it is off. With the transistor off, entry of the main flux intobranch 42 is not longer resisted. The main flux may now enter branch 42and does produce inverse voltage at negligible current across the inputjunction. Had the "virtual electric generator" become a voltage sourcebefore transistor turn off, then a series resistor developing severalvolts and increasing the circuit dissipation several times over that ofthe actual base junction losses would have been required to protect thetransistor during turn-offs. Thus, additional power is saved by thepresent circuit beyond that attributable to efficient operation of thetransistor switch alone.

While the invention is applicable to a variety of ferrite materials, thematerials classified as "soft" ferrites are most satisfactory. Asuitable material in this class is the Stackpole material 24B. Thematerial has a steeper slope and substantial curvature at lower "B"values, but as B increases, both a curvature remains and an absoluteslope remains, though reduced. This property facilitates the function ofthe transistor input junction in sustaining current generator action andprecluding premature voltage generator action. As the second branch 40is saturating under the growing main flux, by which is meant acontinuing reduction in slope of the B/H characteristic is taking place,the two reluctance dependent terms of expression 14 (defining the basecurrent drive) become increasingly negative. This suggests that at somepoint on the slope of the B/H characteristic, the base drive will becomezero, and that beyond this point, the base drive will become negative incorrespondence to observation. To achieve an initial regenerativeaction, a certain slope of the B/H curve is assumed in expression 14.For the final degenerative action, the slope must change enough toincrease the two reluctance dependent terms to the point where the fullexpression becomes negative corresponding to a reversal in drive. Thedesired change in drive from regenerative to degenerative may thereforetake place without precise dependence on the manner in which the B/Hcurve goes from a steeper slope to a less steep slope. However, as notedearlier, the desired current transformer action is facilitated by theassumption that the B/H curve still retains appreciable slope and sodoes not present a zero load to the transistor input junction during theinterval that it itself is a source of energy. Observation of theconstancy of the V_(eb) voltage shows that soft ferrites generally havethe required B/H properties necessary for safe and efficient turn off ofconventional junction type power transistors.

In high efficiency circuits, a damping resistor (35, FIG. 1) isgenerally used to damp any ringing which might occur during switchingand prevent accidental retriggering of the transistor. The resistor,which is connected in shunt across the base junction, prevents highfrequency energy coupled through the collector-base capacity to the baseelectrode from causing injection. The resistor can be chosen so that thedissipation is small compared to the base junction dissipation andpreserves circuit efficiency.

Transformer-transistor circuits of the nature herein contemplated havean improved stability of power versus temperature. This is true in bothtriggered and self-oscillating configurations. Both the junctiontransistor and the ferrite of the core are exposed to the sametemperature in conventional packaging. This intimacy is a consequence ofthe desirability to achieve a minimum size total package and is in partdue to the requirement to reduce electro-magnetic interference whichincreases as lead lengths and component separations increase. Withincreasing temperature, the "B max" of the ferrite decreases, reducingthe volt time area of individual conduction periods. Thus, withincreasing temperature and fixed supply voltage, the ferrite actiontends to decrease the individual conduction periods. The voltage intowhich the control winding operates is a diode junction whose voltagedecreases with temperature, which tends to increase the individualconduction periods. These two effects are approximately in a ratio of 3to 2 with the effect of the ferrite dominating, and producing a 3 to 1improvement in power stability.

While the transformer herein described has been illustrated in use intwo kinds of static inverters, it is also applicable to blockingoscillators and other forms of static inverters.

What is claimed as new and desired to be secured by Letters Patent of the United States is: .[.1. A transformer having control winding..].
 4. .[.A transformer as.]. .Iadd.The combination .Iaddend.set forth in claim .[.3.]. .Iadd.20 .Iaddend.whereinthe reluctances of said first and second branch are equal and the reluctances of said third and fourth branches are equal.
 5. .[.A transformer as.]. .Iadd.The combination .Iaddend.as set forth in claim .[.3.]. .Iadd.20 .Iaddend.whereinthe reluctances of said first and second branches are equal and the reluctances of said third and fourth branches are equal and larger than the reluctances of said first and second branches, predisposing said fourth branch in said first diagonal pair to saturate first. .[.6. A transformer as set forth in claim 5 wherein said first aperture is of lesser diameter than said second aperture..].
 7. .[.A transformer as.]. .Iadd.The combination .Iaddend.set forth in claim .[.6.]. .Iadd.20 .Iaddend.whereinthe reluctance of said fifth branch is less than that of said other branches to preclude saturation thereof prior to said first and fourth branches. .[.8. A transformer as set forth in claim 2 wherein said first and second control windings are of a few turns and are closely coupled in the absence of selective saturation to achieve substantial current transformer action..]. .[.9. A transformer as set forth in claim 8 wherein said first control winding has maximum core coupling to said second control winding in the absence of saturation, said coupling being reduced as each branch saturates..].
 10. In combination,(1) a core of substantially linear magnetic material for main flux .[.persing.]. .Iadd.persuing .Iaddend.a closed magnetic path, a pair of apertures in a localized region of the core, a first aperture being arranged before a second aperture along said magnetic path, said first aperture dividing the magnetic path into a first and second branch, and the second aperture dividing the magnetic path into a third and fourth branch, with a fifth branch being formed between said apertures, the first and fourth branches forming a first diagonal pair of branches and the second and third branches forming a second diagonal pair of branches, (2) a primary winding encircling said magnetic core for generating a main flux in said closed magnetic path when an alternating voltage is applied thereto, (3) a first control winding encircling said fifth branch for generating a circulating flux forming two counterrotating loops, one around each aperture, when an alternating current is supplied thereto, the flux in said two loops combining additively in said fifth branch, .[.(4) means for energizing.]. said primary winding and said first control winding with alternating quantities having a .Iadd.suitable .Iaddend.fixed relative phase .[.so that.]..Iadd., generating a combined flux in which .Iaddend.the flux in one of the loops combines .Iadd.additively .Iaddend.with said main flux .[.additively.]. in each of said first diagonal pair of branches and subtractively in each of said second diagonal pair of branches, predisposing a branch in said first diagonal pair to saturate first as the energization increases, said energization being increased until saturation is approached, .[.and .]. .[.(5).]. (4) a second control winding encircling said fifth branch for deriving an electrical quantity whose sign reverses as a function of the magnetic state of said core.Iadd., and (5) a junction transistor switching device for applying alternating current to said primary winding and to said first control winding, said device exhibiting appreciable stored charge, and (6) means coupling said second control winding across the input junction of said transistor for applying a current to said switching device in a sense aiding normal conduction in the absence of saturation, and in a sense inhibiting conduction when saturation occurs.Iaddend.. .[.11. The combination as set forth in claim 10 having in addition thereto a resistive load coupled to said second control winding, saturation of said first diagonal pair of branches tending to force main flux into said fifth region and reverse the polarity of the electrical quantity coupled to said resistive load..]. .[.12. In combination with a transformer as in claim 10,a junction transistor switching device for applying alternating current to said primary winding and to said first control winding, said device exhibiting appreciable stored charge, and means coupling said second control winding across the input junction of said transistor for applying a current to said switching device in a sense aiding normal conduction in the absence of saturation, and in a sense inhibiting transistor conduction when saturation occurs..].
 13. The combination set forth in claim .[.12.]. .Iadd.10 .Iaddend.wherein said .[.transformer.]. core material has a B max which decreases with increasing temperature tending to reduce the volt time area of each conduction period while the voltage of the transistor input junction to which the .[.transformer.]. .Iadd.second .Iaddend.control winding is connected decreases with increasing temperature so as to increase each conduction period, said connection reducing the effect of temperature upon the output power.
 14. The combination set forth in claim .[.12.]. .Iadd.13 .Iaddend.whereinsaid coupling means couples said second control winding directly across said input junction to provide a serial path of low resistance to reduce circuit dissipation, said second control winding not appearing as a voltage source until said transistor is substantially non-conductive.
 15. The combination set forth in claim 14 whereinthe reluctances of said first and second branches are substantially equal and the reluctances of said third and fourth branches are equal, said input junction sustanining a constant voltage drop across said second control winding and forcing a constant rate of change of flux in said fifth branch so long as said transistor is conductive, saturation of said first diagonal pair of branches increasing the reluctance coupling said second control winding to said first control winding and causing a reversal in polarity of said applied current until stored charge is removed from said switching device and conduction is terminated.
 16. The combination set forth in claim 14 whereinthe reluctances of said first and second branches are equal and the reluctances of said third and fourth branches are equal and larger than the reluctances of said first and second branches, predisposing said fourth branch to saturate first and said first branch to saturate second, saturation of said fourth branch increasing the reluctance coupling said second control winding to said first control winding and reducing the rate of increase of applied current; saturation of said first branch causing a reversal in slope of said applied current and a reversal in current, said reversals continuing until stored charge is removed from said switching device and conduction terminated. .[.17. A transformer having(1) a core of substantially linear magnetic material having a closed magnetic path, a first pair of apertures in the core in a first localized region of the core, and a second aperture pair in a second localized region of the core, one aperture of each said pair dividing the magnetic path into a first and a second branch and the other aperture of that pair dividing the magnetic path into a third and fourth branch, with a fifth branch being formed between said apertures, the first and fourth branches forming a first diagonal pair of branches and the second and third branches forming a second diagonal pair of branches to form a set of five branches in each region, (2) a pair of primary windings encircling the magnetic path for generating a main flux in said core when current is supplied thereto, (3) a pair of first control windings each encircling the fifth branch of each region for generating a circulating flux forming two counterrotating loops around said apertures when current is supplied thereto, the fluxes in said two loops combining additively in said fifth branch, the flux in one of the loops combining with said main flux additively in each of one diagonal pair of branches and subtractively in each of the other diagonal pair of branches, predisposing a branch in said first diagonal pair of saturate first as energization increases, (4) a pair of second control windings, each encircling the fifth branch of said region for deriving an electrical quantity whose sign reverses as a function of the magnetic state of said core..]. .[.18. A transformer as in claim 17 wherein the apertures of each pair are serially arranged along said magnetic path, placing said fifth branch orthogonal to said main flux to reduce the tendency of the main flux to flow into said fifth branch and to be coupled to said second control winding..]. .Iadd.19. The combination set forth in claim 10 wherein said fixed relative phase is achieved by the connection of said first control winding in series with said primary winding. .Iaddend. .Iadd.20. The combination set forth in claim 19 whereinsaid apertures are serially arranged along said magnetic path placing said fifth branch orthogonal to said main flux, and the product of the reluctances of said first diagonal pair of branches equals the product of the reluctances of said second diagonal pair of branches to reduce the tendency of main flux to be coupled to said second control winding. .Iaddend. 